你所使用的D2类型和你选择的R7 和R8的阻值也对响应时间有影响。TL431CLP的一个测试电路，其中R7 是1kΩ，R8是4.7kΩ，对于瞬态过流的响应时间是550ns。用ZR431CLP更换TL431CLP，其响应时间约为1µs。增加R7 和R8的阻值分别到10和47kΩ，则响应时间为2.8µs。注意到TL431CLP产生较大的阴极电流需要相应阻值较小的R7和R8。
为了设定过压跳闸水平在18V，R3 和 R4必须具备阻值分别为62和10kΩ。测试电路实验得出如下结果：D2采用TL431CLP，电路在17.94V跳闸，D2采用ZR431CLP，跳闸电压为18.01V。依靠Q2的基极-发射极电压，过流检测机制的精度低于过压功能。然而，用一个高端电流检测放大器产生与负载电流成正比的地电流来取代R6和 Q2，过流检测精度将大大提高。这些器件可从Linear技术公司、Maxim、德州仪器公司和Zetex等公司得到。
电路断路器被证明是很有用的应用，例如汽车系统，需要过流检测，以防止错误载荷；还需要过压保护，屏蔽敏感电路受到高能负载瞬变时的影响。除了流过R3 和R4的小电流，以及D2的阴极电流，在正常、非跳闸情况下，对于电源，电路没有电流流出。
英文原文：
Circuit breaker provides overcurrent and precise overvoltage protection
A simple circuit breaker delivers precision overvoltage protection and overcurrent protection.
Anthony H Smith, Scitech, Bedfordshire, England; Edited by Brad Thompson and Fran Granville -- EDN, 6/7/2007
Requiring only a handful of inexpensive components, the circuit breaker in Figure 1 responds to both overcurrent- and overvoltage-fault conditions. At the heart of the circuit, D2, an adjustable, precision, shunt-voltage regulator, provides a voltage reference, comparator, and open-collector output, all integrated into a three-pin package.
Figure 2 shows a simplified view of the ZR431, D1. The voltage appearing at the reference input is compared with the internal voltage reference, VREF, nominally 2.5V. In the off state, when the reference voltage is 0V, the output transistor is off, and the cathode current is less than 0.1 µA. As the reference voltage approaches VREF, the cathode current increases slightly; when the reference voltage exceeds the 2.5V threshold, the device fully switches on, and the cathode voltage falls to approximately 2V. In this condition, the impedance between the cathode and the supply voltage determines the cathode current; the cathode current can range from 50 µA to 100 mA.
Under normal operating conditions, D2’s output transistor is off, and the gate of P-channel MOSFET Q4 goes through R9, such that the MOSFET is fully enhanced, allowing the load current, ILOAD, to flow from the supply voltage, –VS, through R6 into the load. Q2 and current-sense resistor R6 monitor the magnitude of ILOAD, where Q2’s base-emitter voltage, VBE, is ILOAD×R6. For normal values of
ILOAD, VBE is less than the 0.6V necessary to bias Q2 on, such that the transistor has no effect on the voltage at the junction of R3 and R4. Because the input current at D2’s reference input is less than 1 µA, negligible voltage drops across R5, and the reference voltage is effectively equal to the voltage on R4.
In the event of an overload when ILOAD exceeds its maximum permissible value, the increase in voltage across R6 results in sufficient base-emitter voltage to turn on Q2. The voltage on R4 and, hence, the reference voltage now pull up toward VS, causing D2’s cathode voltage to fall to approximately 2V. D2’s output transistor now sinks current through R7 and R8, thus biasing Q3 on. Q4’s gate voltage now effectively clamps to the supply voltage through Q3, and the MOSFET turns off. At the same instant, Q3 sources current into R4 through D1, thereby pulling the voltage on R4 to a diode drop below the supply voltage. Consequently, no load current flows through R6 because Q2, whose base-emitter voltage is now 0V, has turned off. As a result, no load current flows through R6, D2’s output transistor latches on, and the circuit remains in its tripped state in which the load current is 0A. When choosing a value for R6, ensure that Q2’s base-emitter voltage is less than approximately 0.5V at the maximum permissible value of the load current.
As well as responding to overcurrent conditions, the circuit breaker also reacts to an abnormally large value of the supply voltage. When the load current lies within its normal range and Q2 is off, the magnitude of the supply voltage and the values of R3 and R4, which form a potential divider across the supply rails, determine the voltage at the reference input. In the event of an overvoltage at the supply voltage, the voltage on R4 exceeds the 2.5V reference level, and D2’s output transistor turns on. Once again, Q3 turns on, MOSFET Q4 switches off, and the load becomes effectively isolated from the dangerous transient.
The circuit now remains in its tripped state until reset. Under these conditions, Q3 clamps Q4’s gate-source voltage to roughly 0V, thereby protecting the MOSFET itself from excessive gate-source voltages. Ignoring the negligibly small voltage across R5, you can see that the reference voltage is VS×R4/(R3+R4) in volts. Because D2’s output turns on when the reference voltage exceeds 2.5V, you can rearrange the equation as R3=[(VST/2.5)–1]×R4 in ohms, where VST is the required supply-voltage trip level. For example, if R4 has a value of 10 kΩ, a trip voltage of 18V would require R3 to have a value of 62 kΩ. When choosing values for R3 and R4 to set the desired trip voltage, ensure that they are large enough that the potential divider will not excessively load the supply. Similarly, avoid values that could result in errors due to the reference-input current.
When you first apply power to the circuit, you’ll find that capacitive, bulb-filament, motor, and similar loads having large inrush current can trip the circuit breaker, even though their normal, steady-state operating current is below the trip level that R6 sets. One way to eliminate this problem is to add capacitor C2, which slows the rate of change of the voltage at the reference input. However, although simple, this approach has a serious disadvantage in that it slows the circuit’s response time to a genuine overcurrent-fault condition.